Speed sensing for the third harmonic stator voltage signal

ABSTRACT

A method and system for sensing the rotor speed of a brushless permanent magnet motor wherein a signal containing isolated third harmonic components of the flux density is acquired, and the absolute values of the maximas of the signal are measured, the last measured absolute maxima representing the current rotor speed of the motor.

BACKGROUND OF THE INVENTION

The present invention relates brushless permanent magnet motors. Morespecifically, the invention relates to interactive sensing methods andapparatus employing the third harmonic component of the stator voltagesignals of such motors to control operation of such motors.

A brushless permanent magnet (BPM) motor, also referred to as apermanent magnet brushless direct current (PMBDC) motor, a permanentmagnet self-synchronous motor or an electronically commutated motor, isa type of motor that comprises a multi full pitch concentrated windingstator configuration with different possibilities for the number ofphases and poles, and a rotor that has permanent magnets mounted in amagnetic structure attached to the motor shaft. The magnets can beeither mounted on the surface of the rotor structure (surface mounted orinset permanent magnet motor) or inside it (buried or interior permanentmagnet motor). The BPM motor is driven or operated by controlledapplication of current signals to the stator windings.

During operation, the rotor magnets produce an air gap flux densitydistribution that is a function of the type of their magnetizationcharacteristic and fabrication process. When the magnets are magnetizedaxially, a trapezoidal air gap flux density is produced. When thismagnetization is parallel to the magnet main axis a sinusoidal air gapflux density is generated. Because the main flux is produced by magnetsthat do not carry currents, motor losses occur that are restricted tothe copper and iron losses in the stator and to iron loss in the rotor.Hence, a BPM motor is suitable for applications where high efficiency isa concern.

Due to their high efficiency and relative control simplicity, BPM motorsare becoming preferred in appliance applications such as compressors,fans, pumps, and washers. Yet, in order to operate a BPM motoradequately, information about the position of the rotor is necessary.This information is used to define stator currents which are applied byan inverter so that the flux produced by these currents is always keptin quadrature with the rotor flux. This allows a complete decouplingbetween rotor flux and stator current vectors, and the result is a motorthat has speed and torque proportional to the voltage and currentamplitude, respectively, similarly to a direct current (DC) motor.

It is possible to sense the back electromotive force (EMF) of a motor toestimate the position of the rotor. However, the back EMF signal cyclesonly once per revolution of the rotor producing only two zero crossingsper cycle and thus is not entirely suitable for controlling statorcurrents that must be defined three times more often during a revolutionfor a three-phase motor because the rotor position can only be estimatedtwice per revolution. Moreover, back EMF signals can be noisy, andfilters therefor can introduce delay.

The general practice is to calibrate operation of a BPM motor forefficiency at one speed. Usually this is accomplished by detecting zerocrossings of the back EMF signal and then gating current applicationbased on preselected delays, the delays accommodated efficient operationat one speed. But at other speeds, the delays are not entirely suitable.Thus, the BPM motor operates inefficiently at other speeds.

In FIG. 1 there is illustrated the idealized air gap flux densitydistribution in a BPM motor with magnets radially magnetized. It isillustrated that the resultant trapezoidal air gap flux density has adominant third harmonic component that links the stator windingsinducing a third harmonic voltage component in each one of the phases.Other high frequency components such as 5^(th), 7^(th) and 11^(th)harmonics, and a switching frequency with its side bands, are alsopresent in the air gap flux, but they are weak relative to the thirdharmonic and thus the third harmonic is the dominant component.

In a three-phase system, all third harmonic voltage components are inphase, forming a zero sequence set. A third harmonic voltage componentis induced in the stator phases and corresponds exactly to the air gapthird harmonic component because no third harmonic currents cancirculate in star connected stator windings.

It can be appreciated that a summation of the three stator phasevoltages results in the elimination of all polyphase components like thefundamental and characteristic harmonics. Only the third harmonic, andother triplens together with the PWM switching frequency and its sidebands will be present in the adder output signal, the third harmonicbeing the dominant component. The result is a signal that can be used toidentify rotor position that cycles three times per rotor revolution,and this provides more accurate rotor positional information than doesonly a back EMF signal.

Further background information regarding BPM motors and means andmethods for obtaining the third harmonic signal are described in thefollowing United States patents, the disclosures of which areincorporated herein by reference:

4,481,440

4,959,596

4,296,362

4,585,982

4,585,983

4,641,066

5,023,924

4,980,617

4,912,378

4,922,169

U.S. Pat. Nos. 4,912,378 and 4,641,066, in particular, provide excellentbackground discussions.

One concern with the summation of the stator phase voltages as describedabove, is that access to the neutral point connection or node of thestator is necessary. For this purpose, a wire connection to the neutralnode, and although easy to install in the majority of applications, itcan, in some cases, represent extra cost or inconvenience to theinstallation.

Another problem can arise when a BPM motor is operated at high torque orhigh speeds if back EMF sensing is needed by the motor. At high torqueor high speeds, the back EMF no longer is available due to blanking outby the commutation of the invertor.

SUMMARY OF THE INVENTION

A number of inventions are described herein.

The invention of this application provides an apparatus and method forsensing the speed of a BPM motor utilizing the third harmonic componentof the stator voltages.

In an embodiment of this invention, the last measured absolute value ofthe third harmonic component signal is taken as the measure of thecurrent speed of the BPM motor.

Another invention provides a system for controlling a BPM motorutilizing the third harmonic component of the stator phase voltages.

In an embodiment of this invention, a signal including the thirdharmonic component of the stator voltages is filtered to isolate thethird harmonic component and then integrated to produce a time integralthereof. Zero crossings of the time integral signal and the back EMF forone phase are detected and the stator currents are synchronized with themotor and applied as necessary depending on these zero crossings.

Another invention provides an arrangement for obtaining the thirdharmonic component of the stator voltages wherein access to the statorneutral point is unnecessary.

In an embodiment of this invention, a star network of resistors iselectrically coupled to the stator phase nodes of the invertor used todrive a BPM motor such that each phase has a resistor coupled betweenits phase node and an artificial neutral node provided separately andapart from, but in correspondence with, a stator winding neutral node,and the third harmonic component is obtained from across the artificialneutral point of the resistor network, and a reference node providedelectrically between positive and negative nodes of the inverter, or theinverter ground node.

Another invention provides a method for controlling a BPM motor at highspeeds.

In an embodiment of this invention, current to one phase is turned offfor several cycles of the motor and zero crossings of the phase voltageare detected during this period. Thereafter, current application isresumed in synchronization with motor operation.

This invention also provides an advantage over the conventional methodof measuring the internal motor voltages (or back EMF) that it is notsensitive to phase delays introduced by filters operation of a BPM motorat high speeds, when the back EMF sensing method fails.

The inventions provide means to measure the rotor flux position based onthe third harmonic voltage component of the stator phase voltages and touse this signal to generate the correct stator currents that arerequired by the motor to operate in an electronically commutated mode.

The inventions also provide a scheme to operate a BPM motor at highspeeds when the back EMF is not accessible from the stator terminalvoltages.

These and other features of the inventions are discussed in greaterdetail below in the following detailed description of the presentlypreferred embodiments with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a signal graph useful for explaining the idealizedair gap flux density for a BPM motor in relation to the third harmoniccomponent of the stator phase voltages.

FIG. 2 illustrates an inverter arrangement useful for applying statorcurrents to a three-phase BPM motor.

FIGS. 3A-3F illustrates a signal graph useful for explaining therelationship between the back EMF for one phase, third harmoniccomponent of the stator phase voltages, and inverter currents for a BPMmotor.

FIG. 4 illustrates an arrangement for operating a BPM motor.

FIG. 5 illustrates another inverter arrangement that can be used todrive a three-phase BPM motor.

FIG. 6 illustrates an arrangement for operating a BPM motor whereinaccess to the stator neutral node is not needed to obtain the thirdharmonic component of the stator phase voltages.

FIG. 7 illustrates another arrangement for operating a BPM motor whereinaccess to the stator neutral node is not needed to obtain the thirdharmonic component of the stator phase voltages.

FIG. 8 illustrates a microcomputer arrangement for controlling a BPMmotor.

FIG. 9 illustrates in greater detail an analog interface in thearrangement of FIG. 7.

FIGS. 10A-10O illustrates a timing diagram useful for explainingoperation of a BPM motor.

FIGS. 11A-11E illustrate an algorithm for a computer program to controlapplication or commutation of stator currents to a BPM motor.

FIGS. 12A and 12B illustrate time/frequency and duty cycle profiles,respectively, during implementation of the algorithm of FIGS. 11A-11E.

FIGS. 13A-13D illustrates a signal graph useful for explaining a methodfor sensing running speed of a BPM motor.

FIGS. 14A and 14B illustrates a flow chart of an algorithm for acomputer program for determining the speed of a BPM motor.

FIG. 15 illustrates a first application of the inventions, wherein thethird harmonic stator-voltage is used to estimate motor, and compressorspeeds in a refrigeration system.

FIG. 16 illustrates a second application of the inventions, wherein thethird harmonic stator voltage is used to estimate motor and drum speedsin a belt driven washing machine.

FIG. 17 illustrates a third application of the inventions, wherein thethird harmonic stator voltage is used for speed control of a BPM motorin a laundry system.

DETAILED DESCRIPTION OF THE PRESENTLY PREFERRED EMBODIMENTS

As mentioned above, a BPM motor is operated by controlled application orcommutation of currents to the stator windings of the motor. This taskis undertaken by a switching device known as an inverter, in conjunctionwith a suitable controller therefor.

FIG. 2 illustrates an inverter 20 configured to apply various currentsto a three-phase BPM motor 21. As can be seen, the inverter 20 includesa direct current signal v_(I) input across an electrical positive railor node 22 and an electrical negative or ground rail or node 24, alsolabeled g.

The invertor 20 includes three pairs of switches 32, 34 and 36 coupledacross the nodes 22 and 24 to provide the square wave signal generationnecessary to operate or drive the BPM motor. Each pair of switches, 32,34 and 36 is associated with one phase of the BPM motor 21. In thisexample, switch pair 32 is associated with a phase a, switch pair 34 isassociated with a phase b and switch pair 36 is associated with a phasec.

As illustrated, each switch pair comprises a pair of switches coupled inseries across the nodes 22 and 24. Coupled across each switch in theknown manner is a diode. For the purposes of this example, switch pair32 includes switches S₁ and S₄. Switch pair 34 includes switches S₃ andS₆, and switch pair 36 includes switches S₅ and S₂. Between the switchesof each switch pair is a circuit node giving rise to one of the phasesa, b or c, to which the stator winding of the BPM motor are connected.Between switches S₁ and S₄ is a node 40 associated with phase a. Betweenswitches S₃ and S₆ is a node 42 associated with phase b. Betweenswitches S₂ and S₅ is a node 44 associated with phase c.

As mentioned above, each stator winding of the BPM motor is coupled toone of the phase nodes 40, 41 or 44. A winding 46 is coupled to node 40of phase a, a winding 48 is coupled to node 42 of phase b, and a winding50 is coupled to node 44 of phase c.

As can be appreciated, a back EMF voltage develops across each ofwindings 46, 48 and 50 during operation of the BPM motor. These back EMFvoltages are represented in FIG. 2 by the references E_(a), E_(b) andE_(c). The windings 46, 48 and 50 are connected in star configurationand have a common neutral node s.

FIGS. 3A-3F illustrates various signals relating to operation of athree-phase BPM motor with the inverter 20. FIG. 3 illustrates back EMFsignal E_(a), a third harmonic component signal v₃, a time integralsignal ∫v₃ dt which is the time integral of the third harmonic componentsignal v₃, and stator winding currents i_(a), i_(b) and i_(c) for thethree phases a, b and c, respectively.

It can be appreciated from FIGS. 3A-3F that a transition in the currentsi_(a), i_(b) and i_(c) exists each time the third harmonic componentsignal v₃ reaches a maximum and, consequently, each time the timeintegral signal ∫v₃ dt crosses zero. Yet detection of the positive zerocrossings of the back EMF voltage for one phase is necessary in order toimplement a control algorithm that can command application of theillustrated stator currents i_(a), i_(b) and i_(c).

As described in greater detail below, a control algorithm can beimplemented that employs the signal information set forth in FIGS. 3A-3Fto define the correct stator currents i_(a), i_(b) and i_(c). In thisregard, an algorithm can be implemented that upon detection of a zerocrossing of the back EMF signal E_(a) (for example, at time t₁) waitsfor the next zero crossing of the time integral signal ∫v₃ dt (in thisexample at time t₂) in order to turn the current i_(a) on for phase aand turn off current i_(c) in phase c. At the next zero crossing of thetime integral signal (in this example at time t₃), the control algorithmturns off the current i_(b) in phase b and turns on the current i_(c) inphase c. This procedure for the turn on and turn off of the currentsthen would continue until one cycle is complete, i.e., at the next zerocrossing of the back EMF signal E_(a). It can be appreciated that thedetection of the zero crossings of the back EMF signal E_(a) for phase ais possible because the phase current i_(a) at the relevant instances iszero, and consequently the terminal phase voltage at the node 40 is thesignal E_(a).

It also can be appreciated that although both the third harmoniccomponent signal v₃ and its time integral ∫v₃ dt cycle three timesduring a motor revolution, and both could be used to controlapplications of the stator currents, it is easier to detect the zerocrossings of the time integral signal ∫v₃ dt than to detect the maximasof the third harmonic component signal v₃. Therefore, the signal ∫v₃ dtis preferred for controlling application of stator currents.

The foregoing technique approach generally requires access to the common(neutral) node s of the stator windings 46, 48 and 50. It also requiresa stator winding pole pitch different than 2/3, otherwise the thirdharmonic flux component does not link the stator windings and the thirdharmonic component is not sufficiently induced in the phases. Yet, animportant advantage of this technique, besides its simplicity, is itslow susceptibility to noise. The result of the summation of the threephase voltages from nodes 40, 42 and 44 contains the third harmonicsignal and high frequency zero sequencing components that can beeliminated by filtering. This filtering action need not necessarilyintroduce a significative phase change in the third harmonic signalwhich would be well below the filter cut off frequency, given a low passfilter. This technique also allows for operation over a wider range offrequencies than the most common technique in use today which merelydetects zero crossings of the back EMF voltages.

As will be discussed in further detail below, if the detection of thezero crossing for the back EMF voltage signal E_(a) is not available forany reason, for example, when the BPM motor operates at high torque orfrequencies, an alternative control scheme can be used. In this scheme,any zero crossing of the integrated signal with a positive slope couldbe detected and the phase a current i_(a) be commanded to a positivevalue for a complete cycle of the time integral signal ∫v₃ dt. At thatsame instant, the current i_(b) in phase b would be commanded to anegative value and current i_(c) in phase c commanded to turn off.Thereafter, the currents would be defined to follow the periodicsix-step waveform.

As additionally will be discussed in further detail below, the speed ofthe BPM motor can be detected for this condition and stored for a futurecomparison. The control algorithm then would select another positiveslope zero crossing of the time integral signal ∫v₃ dt, but now oneperiod of the third harmonic signal away from the first zero crossingselected. The current commands then would be defined, as before, for thefirst point selected and the motor speed sensed and stored. Finally, thealgorithm would select the last option of a zero crossing with positiveslope possible for a complete period of the back EMF signal. At this newpoint, the current commands would be generated and the motor speedmeasured as before. The three values of speed stored during this processwould be compared, and if a constant load is assumed, the crossing pointthat allows maximum motor speed would be selected since this would bethe point that the commanded currents are in phase with the back EMFsignal E_(a) (or in quadrature with the rotor flux) and the motor torqueis maximum.

For instance, suppose that the positive slope crossing indicated at timet₄ in FIGS. 3A-3F is selected first by the control algorithm. At thatinstant the phase a current i_(a) would be turned on, current i_(b) inphase b would set to a negative value, and current i_(c) in phase cturned off. The speed of the motor then would be measured and stored ata memory location of a microprocessor used to implement the controlalgorithm. The next two points in time that the controller would selectare the times t₅ and t₂ indicated in the same figure. The motor speedfor these three possibilities will be maximum at time t₂ where thetorque produced by the motor is maximum. The control algorithm thenwould lock at that position keeping the synchronism with the timeintegral signal ∫v₃ dt. This search for the proper current command wouldtake at most three cycles to be completed, and once the correct currentfiring position is achieved, no need for this process to be repeatedexists unless synchronism with the third harmonic signal is lost.

FIG. 4 illustrates a BPM motor drive system in which can be implementedthe method described above. In FIG. 4, alternating current v_(ac) isfull wave rectified by an input rectifier 51 and then filtered by anappropriate boost converter 52. The resulting DC current then is appliedby an inverter bridge 54 appropriately coupled to a BPM motor 56 asdescribed above. The BPM motor 56 is supplied by the inverter bridge 54with current waveforms like those illustrated in FIG. 3 and describedabove.

As also illustrated, the motor phase voltages are summed by a summingdevice or unit 58 appropriately electrically coupled to each statorwinding and to the stator neutral node s to generate the third harmoniccomponent containing signal v₃. The resulting signal v₃ is basically athird harmonic voltage component that then is filtered and integrated bya low pass filter and integrator circuit or unit 60. The low pass filterremoves any DC signal that could bias the integration.

The detection of the zero crossings for the time integral signal ∫v₃ dtand the back EMF voltage E_(a) of phase a of the stator is performed bya zero crossing detection circuit or section 62. This information isinput into a microprocessor 64 that by means of a simple rotor positiongenerator algorithm 72 described below defines the correct rotorposition reference for the applied stator currents. The speed of themotor 56 can also be obtained from the third harmonic voltage signal (asdescribed later). Thus, a speed regulator unit 65 defines a referencevalue for the stator currents from a comparison of a command speedsignal 66 with an actual motor speed signal 68, the latter being derivedfrom a software rotor speed calculation routine 69 (described below). Acurrent regulator unit or routine 70 receives information from the rotorposition generator 72 and generates a pulse width modulated (PWM) signalapplied to the inverter 54 via suitable drive circuits 74.

Also illustrated in FIG. 4 are protection circuits or unit 75 used toprotect the drive circuits 74 from over currents and the like.

FIG. 5 illustrates another arrangement by which the inverter 20 can beused to drive the BPM motor 21. For the most part, the arrangement ofFIG. 5 is the same as that of FIG. 2, and thus like reference alphanumeric characters are used to denote like parts. However, unlike thearrangement of FIG. 2, the arrangement of FIG. 5 includes a star network82 of resistors 84 having the same resistance value R connected betweenthe nodes 40, 42 and 44 and a common artificial neutral point n. Notethat the neutral stator point is labeled s and the phases of the statorwindings 46, 48 and 50 also are labeled a, b, and c for computationalpurposes. Note also that the neutral node n is provided separately andapart from, but in correspondence with, the neutral node s. Thefollowing voltage equations can then be derived:

    v.sub.as +v.sub.sn +v.sub.na =0                            (1)

    v.sub.bs +v.sub.sn +v.sub.nb =0                            (2)

    v.sub.cs +v.sub.sn +v.sub.nc =0                            (3)

When the three equations above are added up, the result becomes:

    (v.sub.as +v.sub.bs +v.sub.cs)+3v.sub.sn +(v.sub.na +v.sub.nb +v.sub.nc)=0(4)

As discussed above, the summation of the stator phase voltages resultsin an elimination of the polyphase components, leaving the thirdharmonic component plus high order frequency terms so that one canwrite:

    v.sub.as +v.sub.bs +v.sub.cs =3v.sub.s3 +v.sub.highfreq.   (5)

where v_(s3) represents the total third harmonic voltage component for agiven phase and 3v_(s3) represents the total third harmonic componentfor all phases, which is 3 times the individual components for eachphase since they are all in phase. The signal 3v_(s3) is the same as thesignal v₃ herein.

Assuming that the resistors 84 have identical values R and because theyare connected in star, the summation of the currents i_(ar), i_(br) andi_(cr) therethrough is zero. Hence, the following equation results:

    v.sub.na +v.sub.nb +v.sub.nc =R(i.sub.ar +i.sub.br +i.sub.cr)=0(6)

and consequently the voltage between the artificial neutral node n andthe actual stator neutral node s contains the third harmonic voltageplus the high frequency terms,

    3v.sub.s3 +v.sub.highfreq.)+3v.sub.sn =0                   (7)

or,

    v.sub.ns =v.sub.s3 +1/3v.sub.highfreq.                     (8)

It is interesting to note that the third harmonic voltage thus can beobtained directly from the voltage across the two neutral nodes s and nand no electronic summation means is necessary to add the three-phasevoltages as described above and in the U.S. patents mentioned above.Furthermore, the problem of needing to access the stator neutral node swith a fourth wire is also solved as explained next.

In that regard, recalling that the electrical negative or ground rail ornode 24 of the dc bus also is labelled as g, one can write the followingvoltage loop equations:

    v.sub.ag +v.sub.gn +v.sub.ns +v.sub.sa =0                  (9)

    v.sub.bg +v.sub.gn +v.sub.ns +v.sub.sb =0                  (10)

    v.sub.cg +v.sub.gn +v.sub.ns +v.sub.sc =0                  (11)

Assuming that the motor is operating in a commutated mode, where onlytwo switches of the inverter bridge are conducting at any time andassuming that switches S₁ and S₂ are in conduction, one has: ##EQU1##with v_(I) representing the dc voltage input to the inverter bridge 80.After substituting these results in equations (9) to (11) and using theresult from equation (8), one obtains: ##EQU2##

This last equation shows that the voltage across the artificial neutralnode n and the electrical negative or ground rail or node g of the dcbus contains the third harmonic signal and the high frequency termsadded to a dc level given as v_(I) /2. The same happens for othercombinations of switches in conduction, like S₂ -S₃, S₃ -S₄, S₄ -S₅, S₅-S₆, and S₆ -S₁, that correspond to a six-step inverter operationsequence. For all of these switching combinations, th voltage v_(gn) isgiven as in equation (13).

What happens to the voltage v_(gn) when a pulse width modulating (PWM)switching technique is applied to the inverter can also be considered.First, in the case of a 4-quadrant PWM, assuming that S₁ and S₂ are inconduction, the voltage v_(gn) is exactly as in equation (13). When S₁and S₂ are turned off, because of the inductive nature of the loadcurrent, the diodes across switches S₄ and S₅ go into conduction sothat, ##EQU3## and again from equations (11) and (8), ##EQU4## This isexactly like equation (13), showing that when 4-quadrant PWM is used,the third harmonic signal is easily obtained from the voltage v_(gn) bymeans of filtering out the dc level and the high frequency components.

When a 2-quadrant PWM is utilized, after S₁ and S₂ being in conduction,for instance, after S₂ is turned off and the diode across S₅ turns on,the equation

    v.sub.ag =v.sub.I, v.sub.cg =v.sub.I, v.sub.sc =0, and v.sub.sa =0(16)

results from equations (11) and (8), and then the following equation isobtained,

    v.sub.gn =-v.sub.ns -v.sub.I =-v.sub.s3 -1/3v.sub.highfreq. -v.sub.I(17)

In this case, an ac, instead of a dc level varying from v_(I) /2 tov_(I) at the switching frequency rate is also present in the voltagev_(gn). Again, the third harmonic signal can be easily obtained byfiltering v_(gn) with a low pass filter only.

Another switching possibility concerns the 180 degree conduction anglefor the inverter switches. This type of switching is frequently used inac motor drives like permanent magnet/alternating current (PM/AC)motors, induction motors, synchronous motors and so on. In this case,three switches are conducting at any given time. Thus assuming that S₁,S₂, and S₃ are in conduction, one has,

    v.sub.ag =v.sub.I, v.sub.bg =v.sub.I, v.sub.cg =0, v.sub.sc =2/3v.sub.I, and v.sub.sa =-1/3v.sub.I                                 (18)

and the results in equations (11) and (8) yield,

    v.sub.gn =-v.sub.ns -2/3v.sub.I =-v.sub.s3 -1/3v.sub.highfreq. -2/3v.sub.I( 19)

This last equation (19) shows that the third harmonic signal is alsopresent in the voltage v_(gn).

Also coupled across the rails or nodes 22 and 24 is a resistor network86 comprising two series connected resistors 87 and 88 having equalvalues Rdc and a DC midpoint reference node h between them. As can beappreciated, the DC term, v_(I) /2 in equation (13) can be eliminated ifDC reference node h between resistors 87 and 88 is employed as thereference point instead of the negative rail g of the DC bus. Hence,since v_(hg) =v_(I) /2, the following equation results:

    v.sub.hn =-v.sub.s3 -1/3v.sub.highfreq.                    (20)

During inverter commutation, for example, when a switching sequenceswitches from S₁ -S₂ to S₂ -S₃, three switches are closed, i.e., inconduction, at the same time. In this last example, switches S₂ and S₃and the diode across switch S₁ would be in conduction at the same time.At that time, all inverter terminal voltages can be defined by thefollowing:

    v.sub.ag =v.sub.I, v.sub.bg =v.sub.I and v.sub.cg =0       (21)

Utilizing these constraints in equations (9) to (11), the followingresults:

    v.sub.gn =-v.sub.s3 -1/3v.sub.highfreq. -2/3v.sub.I        (22)

and ##EQU5##

During a commutation in which two switches are connected to the node g,for example during the commutation sequence S₂ -S₃ to S₂ -S₃ -S₄ to S₃-S₄, the voltage v_(ng) becomes: ##EQU6##

It can be appreciated from equations (17) and (18) that an alternatingcurrent component that varies between -v_(I) /6 and v_(I) /6 issuperimposed on the third harmonic component signal v_(s3). Since sixcommutations occur in any given period or cycle of the fundamentalinvertor output voltage, this superimposed component has a frequencythat is three times the fundamental frequency and can be considered as acommutation notch which occurs at the same frequency as the thirdharmonic component signal v₃ when the motor is driven with a six-stepwaveform, i.e., no pulse width modulation. The presence of the thirdharmonic component in the signal v_(hn) is not clear when the motor isPWM driven because a commutation notch of ±v_(I) /6 would be generatedat the PWM frequency.

In any event, the third harmonic signal v₃ can be obtained either fromacross the artificial neutral node n and the stator neutral node s orfrom the artificial neutral node n and the DC bus midpoint referencenode h, despite the switching method used for the inverter. Although afilter is necessary to eliminate the undesired high frequency switchingcomponents and any DC signals, it is now clear that the third harmoniccomponent signal v₃ can be obtained without direct access to the statorneutral node n, thus eliminating the need for a fourth wire connectionto the motor.

FIG. 6 illustrates the main components of a BPM motor drive system forthe implementation of the methods described above. In FIG. 6, a BPMmotor 90 is supplied by an inverter 92 with current waveforms like thosein FIGS. 3A-3F described above. Three identical star or y-connectedresistors 94 are used to derive the artificial neutral node n asdescribed above. The voltage v_(gn) between the negative dc bus rail gand the artificial neutral node n contains the third harmonic signal v₃as described above. The signal v₃ is filtered by a suitable filtercircuit or section 96 to eliminate the high frequency components in thesignal. If other than a 2-quadrant PWM technique is used, a DC levelwill be present in the voltage signal v_(gn) that has to be eliminatedby a low pass filter, which also would be present in the filter circuitor unit 96.

After filtering, the signal v₃ is integrated by integrator unit 98 toproduce the time integral signal ∫v₃ dt. The time integral ∫v₃ dt isinput into a zero crossings detection circuit or unit 100 that detectsthe zero crossings for the time integral of the third harmonic signal.The terminal voltage of phase a is measured and also is filtered by alow pass filter in filter circuit or unit 96. To be sure, preferably thesame filter is not used for both the third harmonic signal and the phasesignal. Instead, the phase signal preferably is filtered by a separatesimple anti-aliasing low pass filter. The zero crossings of the filteredphase voltage v_(a) is also detected by the zero crossing detectioncircuit or unit 100.

The third harmonic integral zero crossing signal is then input into aninterrupt port 102 of a microcomputer 104. Every time the third harmonicintegral signal crosses zero an interrupt service subroutine can beexecuted and depending on the zero crossings of the output signal forthe filtered phase voltage, the proper phase current turn ons and turnoffs can then be commanded.

The speed of the motor 90 can also be obtained form the third harmonicvoltage signal v₃ (as described later). For that purpose, a softwareimplemented rotor/motor speed algorithm 107 generates an actual motorspeed signal 108. A speed regulator unit 109 defines the reference valuefor the stator currents from a comparison of a command speed signal 110with an actual motor speed signal 108. The speed regulator unit 109receives information from a rotor position generator unit 111 andgenerates a current reference signal i_(ref) which is used by a currentregulator unit 112 to generate a PWM signal that in turn is applied tothe inverter 92 via suitable drive circuits 113. Also input into thecurrent regulator 112 is a rotor position signal derived from a rotorposition generator unit 111 based on the zero crossings of the signalsv_(a) and ∫v₃ dt.

FIG. 7 illustrates another BPM motor drive system that differs slightlyfrom that of FIG. 6. In the system of FIG. 7, like components areidentified with reference alphanumeric characters like those in FIG. 6.

In FIG. 7, the third harmonic signal is taken from between the DCmidpoint reference node h between resistors 116 and 117, and theartificial neutral node n of the star resistor network 94. As can beappreciated, in the system of FIG. 7, no integrator is employed.Instead, the third harmonic component containing signal is filtered andthen the filtered third harmonic component signal v₃ and phase voltagesignal vas are directly input into the zero crossing detection circuit100, as the time integral is not necessary as described above. Signalsivas and iv₃, described above, then are communicated to the rotorposition and rotor speed units 114 and 107, respectively. In all otherrespects, the systems of FIGS. 6 and 7 are alike.

FIG. 8 illustrates how a BPM motor 120 can be controlled with amicrocomputer 122 employing either the three or four wire connectionscheme described above. As illustrated, an analog interface 124 iscoupled to three lines for the three phases a, b and c extending aninverter 126 and the BPM motor 120 to a digital input/output section 128of the microcomputer 122. The interface 124 also can be coupled to thestator neutral node s, if appropriate, depending on which of the 3 or 4wire connection schemes described above is being employed. Themicrocomputer 122 applies signals to the inverter 126 to drive same asis appropriate to drive the motor 120 by gating the stator windingcurrents as described above. As illustrated, the microcomputer 122, ofcourse, can accept control signals 130 so that particular operation ofthe motor 120 can be selected.

In FIG. 9, there is illustrated in greater detail a portion of theanalog interface 124 of FIG. 8. As illustrated, the interface 124accepts as inputs a third harmonic signal v₃ and a back EMF signal vasvia parallel processing paths. The phase terminal voltage signal v_(as)is first processed through an anti-aliasing low phase filter 132 andthen through an analog/digital converter 134, i.e., a square wavegenerator. The resulting signal is a squared and filtered signali_(vas), which when the phase currents i_(a) is off, is the same as afiltered and squared back EMF signal E_(a).

In parallel, the third harmonic signal v₃ is first processed through ananti-aliasing low pass filter 136, then through a high pass filter 138,and then through an analog/digital converter or square wave generator140, to produce a squared and filtered third harmonic signal iv₃. Thefilters 136 and 138, of course, can comprise a band pass filter.

FIGS. 10A-10O show a simplified time diagram used to describe theimplementation of the proposed methods via a microcomputer. FIG. 7depicts three motor phase voltages, v_(as), v_(bs) and v_(cs) ; phasecurrents, i_(a), i_(b) and i_(c) ; the inverter switching signals S₁ toS₆ ; outputs of the analog interface circuit 124, iv₃ and ivas; and acounter kiv3. The switching signals S₁ to S₆ are defined as in FIG. 2.The signal iv3 ideally is displaced 90° with respect to the thirdharmonic voltage signal for the entire speed range since the low passfilters 132 and 136 preferably are designed with a low cut-offfrequency. Transitions on the signal iv3 correspond to maximum voltage,or rotor flux zero crossings. Therefore, the signal iv3 is actuallydetecting polarity changes for the third harmonic rotor flux component.The signal ivas is likewise phase delayed by 90° with respect to thephase voltage v_(as). If the stator impedance voltage drop is neglected,this signal indicates the change in polarity of the fundamentalcomponent of the air gap flux, which is close to the rotor flux for anon-saturated BPM motor, as explained earlier.

The transitions or zero crossings of the signal iv3 are counted insoftware and the result stored in a counter variable called kiv3. Thesignal ivas is used to reset the software counter kiv3. The signals S1to S6 that are applied to the respective switches or transistors of theinverter bridge are generated according to the state of the counter kiv3as indicated in table 1.

                  TABLE 1                                                         ______________________________________                                        Definition of control signal to the inverter power transistors.               kiv3     S1      S3      S5    S2    S4    S6                                 ______________________________________                                        0        on      off     off   on    off   off                                1        off     on      off   on    off   off                                2        off     on      off   off   on    off                                3        off     off     on    off   on    off                                4        off     off     on    off   off   on                                 5        on      off     off   off   off   on                                 ______________________________________                                    

A flow chart for software developed to control a BPM motor in accordancewith the foregoing is presented in FIGS. 11A-11E. The program executestwo main functions: 1) synchronous starting, and 2) third harmoniccontrol. A main program illustrated in FIG. 11A actually is just aninfinite loop that waits for a timer interrupt to occur. All thesoftware control is provided in an interrupt service subroutine calledINT₋₋ RT illustrated in FIGS. 11B to 11E.

A BPM motor starts from rest as a synchronous motor since the rotor hasto achieve a minimum speed before the third harmonic signal can bedetected. It is known that BPM motors operating in synchronous modepresent an unstable behavior, where current oscillations and high torqueripple can occur. Therefore, it is important to change the motoroperation from synchronous to self-synchronous, or sometimes calledself-commutating, as soon as possible. In this regard, the thirdharmonic control schemes described herein have an advantage over othermethods using only the motor internal voltage (or back EMF) since it iseasier to detect the third harmonic component signal at lower speeds.During tests conducted in the course of investigations of the describedmethods, the third harmonic component signal was acquired afterapproximately two revolutions after a motor was started from rest, at aspeed lower than 50 RPM.

In the next paragraphs, how synchronous starting is implemented in thesoftware illustrated in FIGS. 11A to 11E is described. Signal profilesuseful for explaining such implementation are illustrated in FIGS. 12Ato 12B.

During synchronous starting the applied voltage to frequency ratioconstant is kept constant and, in particular, the frequency is assumedto be an independent variable. It is also assumed that a constant rotoracceleration is desired and a linear time increasing frequency profileis chosen as illustrated in FIG. 12A.

The duty cycle d of the PWM signal applied to the motor is computed fromthe frequency f as:

    d=d.sub.0 +k.sub.d f                                       (25)

where d₀ is a base cycle value and k_(d) is a scaling factor.

The duty cycle profile is shown in FIG. 12B. It is assumed that themaximum frequency is fmax. It is also assumed that the duty cycle is 1.0(or 100%) for this maximum frequency value. The frequency commandincreases linearly with time so that the motor operates with constantacceleration. The frequency f applied to the motor is computed as

    f=k.sub.f t                                                (26)

where t represents time and k_(f) is a scaling factor. The rotorposition RP then can be computed as the integral of frequency.

    RP=∫f dt                                              (27)

In the flow chart of FIGS. 11B-11E, the generated frequency f(n) iscompared to a reference value fref, which is the final frequencyselected for steady state operation. The frequency value fecm is aconstant that defines the frequency value above which the motor iscommanded by the third harmonic signal to operate in self-commutatedmode. While the frequency f(n) is below fecm, the motor operatessynchronously. When in self-synchronous mode, the signal iv3 is read inand the counter kiv3 updated. The switching pattern to the powertransistor is stored in the variable swout. This variable is updatedaccording to the value of the counter kiv3.

The synchronization for the counter kiv3 comes from the detection ofpositive zero crossings of the signal ivas. When a zero crossing of thatsignal occurs, a variable pos₋₋ xing is set to 1. This variable istested in the flow chart section test₋₋ iv3ing and when it is set thecounter kiv3 is reset, and this guarantees the correct synchronismbetween the third harmonic and phase voltage.

If detection of the zero crossing of the signal/vas is not available forany reason, for instance when the motor operates at high speed (or highfrequency), e.g., at 5000 rpm or greater for a 1/4 h.p. motor, althoughwhat is considered high speed will vary depending on the application,alternative control strategies can be used in order to guarantee thecorrect synchronism between the third harmonic and the firing command tothe inverter switches. One way to get the proper synchronism that isdescribed herein is named "advanced turn off". The idea is to turn offone of the motor phases, phase a for instance, so that the currentthrough that phase reaches zero when in high speed operation. The zerocrossing of the phase voltage will then correspond to the zero crossingof the internal motor voltage and the control algorithm can besynchronized with a detection of change in level for the signal ivas.The controller, then, is able to detect the zero crossing (or the statechange of a comparator output of which input is connected to the phase aterminal voltage) and make the decision that phase a is turned on andphase c turned off at the next zero crossing of the signal iv3. Thisprocedure evidently would cause a torque disturbance and because of thatsome care would have to be taken when deciding on the frequency thatthis synchronisms process is utilized and the advance turn off required.If the drive system noise immunity is large, and the third harmonicsignal free of commutation noise, the need for this synchronizingprocess may be necessary every 10000 or so rotor revolutions. In thisinstance, the speed disturbance due to the effects of torque variationscan be greatly reduced, especially at high speeds when this scheme isapplied. It can be appreciated that the number and length of interruptswill vary depending on the application.

FIG. 13 illustrates a timing chart useful for explaining how theoperating speed of a BPM motor can be sensed or identified. In FIGS.13A-13D, the timing relationship between the third harmonic componentsignal v₃, the time integral of that signal ∫v₃ dt, a sampling signal,and a sampled motor speed signal is illustrated. As can be seen, thesampling signal is triggered by detection of zero crossings of the timeintegral signal ∫v₃ dt. Thus, these samples of the motor speed caneasily be taken during one cycle of the motor.

As illustrated in FIG. 13, the third harmonic component signal v₃ issampled at the rate dictated by the sampling signal. The absolute valueof the signal v₃ directly correlates with the motor speed. Thus, thesampled motor speed signal is a DC signal whose value at any giveninstant is directly proportional to the last sampled absolute value ofthe signal v₃.

In FIGS. 14A and 14B there is illustrated in a flow chart, an algorithmfor effecting on a microcomputer the motor speed sensing just described.As illustrated in FIG. 14A, a main program normally runs on themicrocomputer such as that illustrated in FIG. 11A. Then, preferably,upon detection of a zero crossing of the time integral signal, a programinterrupt is caused to occur which calls into operation a motor speedsensing algorithm INT₋₋ SPD, illustrated in FIG. 14B.

As illustrated in FIG. 14B, the interrupt program, the instantaneousamplitude of the signal v₃ first is sensed in step 154. Then the DClevel of the signal v₃ is estimated after low pass filtering in sep 156.Then any DC bias is eliminated in step 158. Then the filtered thirdharmonic component signal v₃ is integrated in step 160. Then zerocrossings of the time integral signal ∫v₃ dt are detected in step 162.If no zero crossing has occurred, then the interrupt program INT₋₋ SPDrecommences at step 154. Otherwise, the peak amplitude of the signal v₃is obtained in step 164.

Finally, in step 166, the absolute value of the sensed peak value of thethird harmonic compound signal v₃ is taken and converted into a positiveDC signal by adjusting the signal by a known factor to produce thesampled motor speed signal described above in connection with FIG. 13.

FIGS. 15, 16 and 17 illustrate in block diagram form at least threeapplications of the foregoing inventions. FIG. 15 illustrates uses ofone or more of the inventions in operating a BPM motor 200 as acompressor motor for a compressor 202 of a refrigerator 204.

As illustrated, a three-phase inverter 206 is controlled by a logiccontroller 208. The logic controller 208 receives signals from a speedcontroller 210 which sets forth the requested motor speed based on asensed compressor speed signal 212 and temperature signals 214 and 216received from a speed sensing algorithm 218 described above andthermostat 220 and user setting 216, respectively. The speed sensingalgorithm 212 in turn receives the various filtered and third harmoniccomponent signals v₃ from an analog interface 218 appropriately coupledto the stator phases of the BPM motor 200 as described above.

FIG. 16 illustrates a somewhat similar arrangement for controllingoperation of a BPM motor 300 for a belt-driven drum 302 of a clothesdrier. The motor 300 is mechanically coupled to the drum 302 by a belt304 in a suitable manner.

As illustrated, a three-phase inverter 306 is used to apply statorcurrents to the motor 300. An analog interface 308 obtains the thirdharmonic signal v₃ by one of the above described methods. A speed memoryalgorithm 310 derives an estimated drum/motor speed signal 312 based onthe algorithm of FIGS. 14A and 14B. The estimated drum/motor speedsignal 312 is received by a controller 314 which also receives controlsignals 3 15 from a console 316. The console 316 in turn receives a drumspeed setting input 318 from a user.

As is also illustrated, the controller 314 generates the above describedgating signals 320 that drive the inverter bridge of the inverter 306.

FIG. 17 illustrates a control scheme for a speed control of a BPM motorin a laundry system that is similar to the control scheme of FIG. 7. Asillustrated, an inverter 400 is coupled to drive a BPM motor 402 and astar network of summing resistors 404 is coupled to the phase nodes ofthe inverter 400 to provide an artificial neutral node n as describedabove. The third harmonic component containing signal is obtained fromacross the midpoint reference node h of the DC bus and the artificialneutral node n of the star network of summing resistors 404.Additionally, the voltage of phase a is obtained.

The foregoing signals are low pass filtered in a filter section 406 toproduce a filtered back EMF signal v_(as) and v₃ are subjected to zerocrossings detection in a zero crossing detector section 408 to producezero crossing signals/vas and iv₃, respectively. The signals ivas, iv₃and v₃ are transmitted as inputs to a microcomputer 4 10 in a mannersimilar to that described above in connection with FIG. 8. However, inthe embodiment of FIG. 17, the signal v₃, not the signal iv₃, isdirected into an analog to digital converter 412, and the resultingdigital signal is directed to a rotor/motor speed algorithm 414 such asthat set forth above in connection with FIGS. 14A and 14B.

Although modifications and changes may be suggested by those skilled inthe art, it is intended that the patent warranted hereon embodies allchanges and modifications as reasonably and properly come within thescope of the contribution of the inventions to the art.

I claim:
 1. A method for determining rotor speed information of abrushless permanent magnet motor, said motor having stator phasewindings relative to which the rotor rotates and in which are induced aback EMF force having a fundamental component and a third harmoniccomponent, said motor being driven by an inverter bridge operativelycoupled thereto, comprising the steps of:generating a first signalincluding the third harmonic component; filtering the first signal andgenerating from the first signal a second signal containingsubstantially only the third harmonic component; integrating the secondsignal and generating a third signal that comprises the integral of thesecond signal; measuring the amplitude of the second signal at instancesdefined by zero crossings of the third signal; and generating a speedmeasurement signal that comprises a value proportional to the absolutevalue of the last measurement of the amplitude of the second signal. 2.The method of claim 1, wherein the step of filtering the first signalcomprises low pass filtering the first signal to remove signalcomponents with frequencies above that of the third harmonic component.3. The method of claim 2, wherein the step of filtering the first signalalso comprises high pass filtering the first signal to remove signalcomponents with frequencies below that of the third harmonic component.4. The method of claim 1, wherein the step of filtering the first signalcomprises high pass filtering the first signal to remove signalcomponents with frequencies above that of the third harmonic component.5. The method of claim 1 further comprising the step of generating aninterrupt in a microcomputer upon occurrence of a zero crossing of thethird signal.
 6. The method of claim 1, wherein the speed measurementsignal comprises a direct current signal whose amplitude is proportionalto the absolute value of the last measurement of the amplitude of thesecond signal.
 7. The method of claim 1, wherein the step of generatingthe first signal includes extracting the first signal from across theground node of the inverter and a common neutral node of the statorwindings.
 8. The method of claim 1, wherein the step of generating thefirst signal comprises extracting a signal from across a node providedmidway between positive and negative rails of the inverter and anartificial neutral node provided separate and apart from, but incorrespondence with, a common neutral node of the stator phase windings.9. The method of claim 1, wherein the step of generating the firstsignal comprises extracting a signal from across a negative rail of theinverter and an artificial neutral node provided separate and apartfrom, but in correspondence with, a common neutral node of the statorwindings.
 10. A system for determining rotor speed information of abrushless permanent magnet motor, said motor having stator phasewindings relative to which the rotor rotates and in which are induced aback EMF force having a fundamental component and a third harmoniccomponent, said motor being driven by an inverter bridge operativelycoupled thereto, comprising:means for acquiring a first signal includingthe third harmonic component; a filter operatively coupled to the meansfor generating the first signal, which filter generates a second signalcontaining substantially only the third harmonic component; anintegrator unit coupled to the filter and configured to generate a thirdsignal signal that comprises the integral of the second signal; a zerocrossings detector coupled to the integrator unit to detect zerocrossings of the third signal and to generate a fourth signal withinformation concerning zero crossings of the third signal; a unitcoupled to both the integrator unit and the filter that measures theamplitude of the second signal at instances defined by zero crossings ofthe third signal; and a unit that generates a speed measurement signalthat has a value proportional to the absolute value of the last measuredamplitude of the second signal.
 11. The system of claim 10, wherein thefilter comprises a low pass filter which removes signal components withfrequencies above that of the third harmonic component.
 12. The systemof claim 11, wherein the filter comprises a high pass filter whichremoves signal components with frequencies below that of the thirdharmonic component.
 13. The method of claim 11, wherein the filtercomprises a high pass filter which removes signal components withfrequencies above that of the third harmonic component.
 14. The systemof claim 10 further comprising a microcomputer having an interrupt portto which the fourth signal is coupled.
 15. The system of claim 10,wherein the speed measurement signal comprises a direct current signalwhose amplitude at a given instant is proportional to the absolute valueof the last measured amplitude of the second signal.
 16. The system ofclaim 10, wherein the means for generating the first signal includeselectrical connections for extracting the first signal from across theground node of the inverter and a common neutral node of the statorwindings.
 17. The system of claim 10, wherein the means for generatingthe first signal comprises an electrical node between positive andnegative power nodes of the inverter, a star network of resistorscoupled to nodes of the inverter and providing an artificial neutralnode separate and apart from, but in correspondence with, a commonneutral node of the stator phase windings and electrical connections forextracting a signal from across the node provided between the positiveand negative power nodes of the inverter and the artificial neutralnode.
 18. The system of claim 10, wherein the means for generating thefirst signal comprises a star network of resistors coupled to nodes ofthe inverter and providing an artificial neutral node separate and apartfrom, but in correspondence with, a common neutral node of the statorwindings, and electrical connections for extracting a signal from acrossa negative power node of the inverter and the artificial neutral node.19. A method for determining rotor speed information of a brushlesspermanent magnet motor, said motor having stator phase windings relativeto which the rotor rotates and in which are induced a back EMF forcehaving a fundamental component and a third harmonic component, saidmotor being driven by an inverter bridge operatively coupled thereto,comprising the steps of:generating a first signal including the thirdharmonic component; low and high pass filtering the first signal andgenerating from the first signal a second signal containingsubstantially only the third harmonic component; integrating the secondsignal and generating a third signal that comprises the integral of thesecond signal; measuring the amplitude of the second signal at instancesdefined by zero crossings of the third signal; and generating a directcurrent speed measurement signal whose amplitude at any given instant isproportional to the absolute value of the last measurement of theamplitude of the second signal.
 20. The method of claim 19, wherein thestep of generating the first signal includes extracting the first signalfrom across the ground node of the inverter and a common neutral node ofthe stator windings.
 21. The method of claim 19, wherein the step ofacquiring the first signal comprises extracting a signal from across anode provided midway between positive and negative rails of the inverterand an artificial neutral node provided separate and apart from, but incorrespondence with, a common neutral node of the stator phase windings.22. The method of claim 19, wherein the step of generating the firstsignal comprises extracting a signal from across a negative rail of theinverter and an artificial neutral node provided separate and apartfrom, but in correspondence with, a common neutral node of the statorwindings.
 23. A system for determining rotor speed information of abrushless permanent magnet motor, said motor having stator phasewindings relative to which the rotor rotates and in which are induced aback EMF force having a fundamental component and a third harmoniccomponent, said motor being driven by an inverter bridge operativelycoupled thereto, comprising:means for acquiring a first signal includingthe third harmonic component; a filter operatively coupled to the meansfor generating the first signal, which filter generates a second signalcontaining substantially only the third harmonic component; anintegrator unit coupled to the filter and configured to generate a thirdsignal signal that comprises the integral of the second signal; a zerocrossings detector coupled to the integrator unit to detect zerocrossings of the third signal and to generate a fourth signal withinformation concerning zero crossings of the third signal; a unitcoupled to both the integrator unit and the filter that measures theamplitude of the second signal at instances defined by zero crossings ofthe third signal; and a unit that generates a speed measurement signalthat has a value proportional to the absolute value of the last measuredamplitude of the second signal.
 24. The system of claim 23, wherein themeans for acquiring the first signal includes electrical connections forextracting the first signal from across the ground node of the inverterand a common neutral node of the stator windings.
 25. The system ofclaim 23, wherein the means for acquiring the first signal comprises anelectrical node between positive and negative power nodes of theinverter, a star network of resistors coupled to nodes of the inverterand providing an artificial neutral node separate and apart from, but incorrespondence with, a common neutral node of the stator phase windingsand electrical connections for extracting a signal from across the nodeprovided between the positive and negative power nodes of the inverterand the artificial neutral node.
 26. The system of claim 23, wherein themeans for acquiring the first signal comprises a star network ofresistors coupled to nodes of the inverter and providing an artificialneutral node separate and apart from, but in correspondence with, acommon neutral node of the stator windings, and electrical connectionsfor extracting a signal from across a negative power node of theinverter and the artificial neutral node.